1. Field of the Invention
The present invention relates to a motor control device for controlling the operation of a motor. The present invention also relates to a motor drive system incorporating such a motor control device.
2. Description of Related Art
There have conventionally been developed motor control devices (position-sensorless control devices) that estimate the rotor position of a motor without the use of a rotor position sensor and controls the motor based on the so estimated rotor position. Sensorless control performed in such motor control devices divides into high-speed sensorless control suitable for a motor in high-speed rotation and low-speed sensorless control suitable for a motor in low-speed rotation or at a standstill.
High-speed sensorless control generally adopts a method relying on the induced voltage (in other words, electromotive force) that appears as the rotor rotates. With this method, however, it is difficult to accurately estimate the position and speed of the rotor in low-speed rotation or at a standstill. For this reason, with the rotor in low-speed rotation or at a standstill, low-speed sensorless control is used that relies on the magnetic salient pole and magnetic saturation of the rotor.
A motor control device is therefore often configured so as to be capable of switching among a plurality of control methods, including one for high-speed sensorless control and another for low-speed sensorless control, so as to be thereby able to achieve stable control in a wide range of speed. Such switching is performed, for example, according to the rotation speed of the rotor.
On the other hand, there has conventionally been disclosed a method whereby a phase generated by a phase determination method for a low-frequency region and a phase generated by a phase determination method for a high-frequency region are subjected to averaging with different weights given at different frequencies, and the resulting phase is estimated to be the phase in a d-q coordinate system.
Now, with reference to FIG. 37, common sensorless control (e.g., high-speed sensorless control) will be described.
FIG. 37 is a block diagram of a conventional motor control device 103. In the configuration shown in FIG. 37, the axes estimated, for control purposes, to correspond to the d-axis and q-axis assumed for vector control of a motor are shown as the γ-axis and δ-axis, respectively. FIG. 39 shows the relationship among the d-axis, q-axis, γ-axis, and δ-axis. In FIG. 39, the symbol Eex represents the voltage vector generally called the extension induction voltage (extended electromotive force).
A current detector 11 detects the U-phase current iu and V-phase current iv in the motor current supplied from a PWM inverter 2 to a salient-pole motor 1. A coordinate converter 12 converts the U-phase current iu and V-phase current iv into a γ-axis current iγ and a δ-axis current iδ. A position/speed estimator 120 (hereinafter referred to simply as “the estimator 120”) estimates and outputs an estimated rotor position θe and an estimated motor speed ωe.
A subtracter 19 subtracts the estimated motor speed ωe fed from the estimator 120 from a specified motor speed value ω*, and outputs the result of the subtraction. Based on the result (ω*−ωe) of the subtraction by the subtracter 19 etc., a speed controller 17 creates a specified δ-axis current value iδ* to be followed by the δ-axis current iδ. Based on the specified δ-axis current value iδ* etc., a magnetic flux controller 116 outputs a specified γ-axis current value iγ* to be followed by the γ-axis current iγ. A current controller 15 outputs a specified γ-axis voltage value vγ* and a specified δ-axis voltage value vδ* such that the current error (iγ*−iγ) and current error (iδ*−iδ) fed from subtracters 13 and 14 both converge to zero.
Based on the estimated rotor position θe fed from the estimator 120, a coordinate converter 18 performs reverse conversion of the specified γ-axis voltage value vγ* and the specified δ-axis voltage value vδ* to create specified three-phase voltage values composed of a specified U-phase voltage value vu*, a specified V-phase voltage value vv*, and a specified W-phase voltage value vw*; the coordinate converter 18 feeds these to the PWM inverter 2. Based on these specified three-phase voltage values (vu*, vv*, and vw*), the PWM inverter 2 creates pulse-width-modulated signals, and supplies the motor 1 with a motor current commensurate with the specified three-phase voltage values to thereby drive the motor 1.
FIG. 38 shows the internal configuration of the estimator 120. The estimator 120 includes an axis error estimator 130, a proportional-plus-integral calculator 131, and an integrator 132. The axis error estimator 130 estimates the axis error Δθ between the d-axis and γ-axis. The axis error estimator 130 calculates the axis error Δθ according to, for example, formula (1) below. Here, Ld and Lq represent the d-axis inductance and q-axis inductance of the motor 1, and Ra represents the motor resistance of the motor 1; s represents the Laplace operator. There have been proposed various methods for estimating the rotor position, of which many use, as an operation parameter, the value of the q-axis inductance of the motor in the estimation formula, as in formula (1) below.
                    Δθ        =                                            tan                              -                1                                      ⁡                          (                                                -                                      E                                          ex                      ⁢                                                                                          ⁢                      γ                                                                                        E                                      ex                    ⁢                                                                                  ⁢                    δ                                                              )                                =                                    tan                              -                1                                      ⁡                          (                                                -                                      (                                                                  v                        γ                        *                                            -                                                                        (                                                                                    R                              a                                                        +                                                                                          L                                d                                                            ⁢                              s                                                                                )                                                ⁢                                                  i                          γ                                                                    +                                                                        ω                          e                                                ⁢                                                  L                          q                                                ⁢                                                  i                          δ                                                                                      )                                                                                        v                    δ                    *                                    -                                                            (                                                                        R                          a                                                +                                                                              L                            d                                                    ⁢                          s                                                                    )                                        ⁢                                          i                      δ                                                        -                                                            ω                      e                                        ⁢                                          L                      q                                        ⁢                                          i                      γ                                                                                  )                                                          (        1        )            
Formula (1) above is a formula for calculating the axis error Δθ. In formula (1), Eexγ and Eexδ represent the γ-axis component and δ-axis component, respectively, of the extension induction voltage Eex.
To achieve a PLL (phase-locked loop), the proportional-plus-integral calculator 131 cooperates with different parts of the motor control device 103 to perform proportional-plus-integral control, and thereby calculates the estimated motor speed ωe such that the axis error Δθ calculated by the axis error estimator 130 converges to zero. The integrator 132 integrates the estimated motor speed ωe outputted from the proportional-plus-integral calculator 131 to calculate the estimated rotor position θe. The estimated motor speed ωe outputted from the proportional-plus-integral calculator 131 and the estimated rotor position θe outputted from the integrator 132 are, as the output values of the estimator 120, fed to those parts of the motor control device 103 which require those values.
With the motor control device 103 configured as described above, the axis error Δθ between the d-axis and γ-axis converges to zero, and this allows stable motor control. Incidentally, while the axis error Δθ is kept equal to zero, the d-axis current id follows the specified γ-axis current value iγ*, and the q-axis current iq follows the specified δ-axis current value iδ*.
The formula for calculating the d-axis current id to achieve maximum torque control exploiting a reluctance torque is well known. In the motor control device 103 configured as described above, to perform maximum torque control, the magnetic flux controller 116 calculates the specified γ-axis current value iγ* according to formula (2) below. Here, Φa represents the armature flux linkage ascribable to the permanent magnet.
                              i          γ          *                =                                            Φ              a                                      2              ⁢                              (                                                      L                    q                                    -                                      L                    d                                                  )                                              -                                                                      Φ                  a                  2                                                  4                  ⁢                                                            (                                                                        L                          q                                                -                                                  L                          d                                                                    )                                        2                                                              +                              i                δ                                  *                  2                                                                                        (        2        )            
Maximum torque control according to formula (2) presupposes that the axis error Δθ is kept equal to zero. On the other hand, calculating the axis error Δθ according to formula (1) above requires, as a calculator parameter (motor parameter), the value of the q-axis inductance Lq. Accordingly, in general, to perform maximum torque control, the actual value of the q-axis inductance Lq of the motor 1 is monitored and, by use of the actual value of the q-axis inductance Lq as it is, the axis error Δθ (and hence the estimated rotor position θe) is calculated.
Moreover, to achieve high-efficiency operation through maximum torque control or the like exploiting a reluctance torque, as will be understood from formula (2) above, a d-axis current id commensurate with the q-axis current iq needs to be passed through the motor. Accordingly, to perform high-efficiency operation by use of the motor control device 103 shown in FIG. 37, the specified γ-axis current value iγ* is calculated according to formula (2).
Moreover, the formula for calculating the specified γ-axis current value iγ* for maximum torque control or the like contains a plurality of motor parameters whose true values are unknown, and thus, if there are errors between the motor parameters (calculation parameters) used to calculate the specified γ-axis current value iγ* and the true motor parameters, it is impossible to perform motor control as desired. For this reason, in the motor control device 103 shown in FIG. 37, adjustments are made beforehand to minimize such errors.
There has conventionally been disclosed, also, the relationship between the errors in the calculation parameters used to estimate the rotor position and the error in the estimated position (axis error).
While the description thus far given with reference to FIGS. 37 to 39 deals with the operation of common sensorless control, conventional motor control devices that use a plurality of control methods switch among different control methods within a single coordinate system so that the rotation axes (d-q axes) to be estimated do not change between before and after a switch. From the perspective of easy adjustment of the operation parameters, reduction of calculation amount, or any other benefit, however, switching within a single coordinate system is not always be the optimum choice.